Software adaptable high performance multicarrier transmission protocol

ABSTRACT

Techniques for reducing peak-to-average power in multicarrier transmitters employ peak cancellation with subcarriers that are impaired by existing channel conditions. The use of Carrier Interferometry (CI) coding further improves the effectiveness of peak reduction. CI coding can also be impressed onto pulse sequences in the time domain, which enhances spectral selection and facilitates peak-power control.

RELATED APPLICATIONS

The present invention claims priority to Provisional Appl. No.60/431,877, filed on Dec. 9, 2002 and Provisional Appl. No. 60/435,439,filed on Dec. 20, 2002, and is a Continuation in Part of patentapplication Ser. No. 09/906,257, filed Jul. 16, 2001 now U.S. Pat. No.6,686,879, which is a non-provisional application of U.S. Pat. Appl.60/219,482, filed on Jul. 19, 2000

FIELD OF THE INVENTION

The present invention relates to multi-carrier communications, and inparticular, to Carrier Interferometry.

BACKGROUND OF THE INVENTION

Multi-carrier modulation can result in a high peak-to-average powerratio (PAPR) of the transmitted signals. The PAPR is defined as theratio of a signal's peak power level to its average power level. In thecase of conventional OFDM modulation, the amplitude of the transmissionis characterized by a substantially Gaussian-shaped probabilitydistribution function. This Gaussian distribution indicates thepossibility that some time-domain samples of the transmission may haveamplitudes that are very high compared to the average sample amplitude.The resulting PAPR is much higher for conventional multi-carrier signalsthan for single-carrier signals. This is due to the probability ofsubstantially constructive interference between the carriers.

The high PAPR for conventional multi-carrier signals imposes significantconstraints on the transmission circuitry, and can greatly complicatethe analog circuitry required for high-fidelity transmission. Forexample, a high PAPR translates into a large dynamic range at the inputsof digital-to-analog converters (DACs) and analog-to-digital converters(ADCs), necessitating a large number of bits for resolution. This cansignificantly increase a receiver's cost and complexity. Highly complexfilters and amplifiers must be employed to handle high PAPR and theincreased resolution. Furthermore, high PAPR results in higher powerconsumption in the transceiver circuits, further increasing the cost ofthe circuits and systems used in the analog front end.

Many prior-art techniques adapted to control the PAPR of multi-carriertransmissions employ clipping to attenuate signal amplitudes that exceeda selected threshold. This results in signal loss and an increase in thebit-error rate (BER). Clipping effectively introduces a cancellationimpulse signal in the time-domain signal. As known in the art, atime-domain impulse corresponds to additive noise across all subchannelsin the frequency domain. Thus clipping effectively reduces thesignal-to-noise ratio for all subchannels in the modulated signal. Thisis not the case for a CI signal. Rather, an impulse constructed fromuser-allocated subcarriers can be applied to the particular CIphase-space(s) of interest without affecting other CI signal phasespaces. This approach either localizes signal degradation (which is aninsignificant problem when channel coding is employed over multiplephase spaces) or slightly reduces bandwidth efficiency, such as byrequiring additional information to be sent to the receiver tocompensate for PAPR-mitigating signal distortions.

A variety of PAPR-reduction methods are disclosed in U.S. Pat. Nos.5,623,513, 5,787,113, 5,768,318, and 5,835,536.

Various approaches have been developed to minimize the number of samplesthat require clipping. According to one class of techniques, datasymbols are coded so that the resulting code words reside in a set oftransmission symbols that reside below a predetermined PAPR threshold.These techniques reduce bandwidth efficiency due to the resulting codingoverhead.

Another well-known approach applies a phase rotation to some of thesubchannels to reduce the probability that a predetermined PAPRthreshold is exceeded. Assuming a low probability that the originalsignal exceeds the PAPR threshold, the probability that both theoriginal signal and the transformed signal exceed the threshold isapproximately the square of the low probability. This is a particularlyuseful technique that can be applied to CI signaling because CI signalsexhibiting high peaks are relatively rare. In a closely relatedtechnique, different data sequences (e.g., a cyclic rotation of thedata) for a given data block may be generated. The sequence having thelowest PAPR is then selected for transmission. These and relatedapproaches can greatly reduce clipping. However, control-signal overheadis required to inform the receiver of changes to the transmissionsignal.

Another approach estimates and corrects the effects of clipping at thereceiver. An estimate of the clipping error is generated at the receiverand used to reconstruct a frequency domain compensation signal to removethe effects of any clipping.

In another PAPR-reduction technique, a normalizer is employed fordetermining a maximum-amplitude value from a plurality of data samples,and then dividing each of the data samples by the maximum amplitudevalue to produce normalized amplitude values. The normalized values maybe provided with non-linear amplification for at least a subset of thevalues.

Another PAPR-reduction technique employs reserved subcarriers to cancelpeaks. A bandwidth-efficient method of reducing the PAPR in DMTtransmissions is disclosed in Gatherer and Polley, “Controlling clippingprobability in DMT transmission”, Proceedings of the Asilomar Conferenceon Signals, Systems, and Computers, (1997), pp. 578-584, the contents ofwhich are herein incorporated by reference. A similar method of reducingPAPR is disclosed in T. Starr, J. M. Cioffi and P. J. Silverman,“Understanding Digital Subscriber Line Technology”, published byPrentice-Hall, 1999, which is incorporated by reference. A predeterminednumber of subcarriers are used to inject symbols that reduce the PAPR ofa DMT signal, and an iterative algorithm teaches which symbols areinjected. However, to reduce the PAPR significantly, up to 20% of thesub-carriers are required to inject the symbols, leaving fewersub-carriers for carrying information. In addition, this method iscomplex. It requires iterative minimization of non-linear functions andcomputation of several fast Fourier transforms.

Each of these PAPR approaches can be used in combination with CI. Thebenefits of such combinations yield a higher performance budget fromwhich advantageous combinations of PAPR, bandwidth efficiency, spectralroll-off, and BER are achieved. The benefits of low initial PAPR in CImodulation enhance the effectiveness of prior-art PAPR-reductiontechniques.

SUMMARY OF THE INVENTION

CI coding can greatly reduce the PAPR of multicarrier signals andenhance other techniques used for PAPR mitigation. The low PAPR enabledby CI coding translates into a smaller dynamic range at the inputs ofdigital-to-analog converters and analog-to-digital converters, enablinga smaller number of bits for resolution. This can significantly reduce areceiver's cost and complexity. Furthermore, low PAPR enables much lowerpower consumption and distortion in the transceiver circuits, furtherreducing the cost of the circuits and systems used in the RF front end.

CI coding spreads data symbols across subcarriers, which combine toproduce superposition pulse waveforms. Sinc-shaped CI pulses have thesame spectral roll-off, but lower PAPR, compared to conventionalmulticarrier modulation. Windowing, or pulse shaping, can further reducethe PAPR of CI signaling by reducing the steep spectral roll-off.Similarly, other PAPR-reduction techniques result in lower spectralefficiency and/or a higher error floor.

PAPR-reduction techniques other than basic CI coding trade off spectralefficiency or BER performance for PAPR benefits. This characterizes onetype of performance budget that is associated with a particulartransmission protocol. At one extreme is multicarrier signaling withhigh spectral efficiency and high PAPR. At the other extreme issingle-carrier modulation characterized by extremely low PAPR and lowbandwidth efficiency. CI coding enables multicarrier modulation toachieve close to the largest performance budget possible. This enablesmulticarrier communications to achieve PAPR benefits (and the relatedperformance metrics) associated with single-carrier modulation.

Improved methods for optimizing the balance of performance metrics in aPAPR-reduction method are disclosed herein. In one aspect of theinvention, the necessary bandwidth required to improve PAPR performanceemploys spectrum that is sub-optimal for data communications. Forexample, peak-reduction signals (e.g., cancellation signals) aretransmitted at sub-carrier frequencies experiencing deep fading orstrong interference. These sub-optimal subcarriers are typicallyunloaded subcarriers (i.e., no data is transmitted on the sub-optimalsubcarriers). Thus, the loss of bandwidth efficiency due to PAPRmitigation is advantageously positioned to coincide with a reduction inbandwidth efficiency due to channel impairments.

Another aspect of the invention exploits a novel transceiver design thatmitigates the effects of a high-PAPR transmission. In particular, ahigh-PAPR signal (e.g., a multicarrier or amplitude-shift key signal) issynthesized by generating low-PAPR signal components that are amplifiedprior to combining. CI signaling allows multiple sub-carrier componentsto be combined into low-PAPR signals prior to amplification, thussubstantially reducing the number of power amplifiers needed.

In another aspect of the invention, a multicarrier signal is generatedby impressing one or more CI codes onto a sequence of pulse waveforms.This facilitates PAPR control prior to modulation with data.

Other objects and advantages of the present invention will be apparentto those of ordinary skill in the art having reference to the followingspecification together with its drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

The present invention is described with reference to the accompanyingdrawings:

FIG. 1A illustrates a software-defined transmitter based on a CI coder.Different CI code vectors are used to synthesize the PHY-layerparameters of a variety of transmission protocols.

FIG. 1B illustrates a software-defined receiver that employs a CIdecoder.

FIG. 2 illustrates a CI transmitter and a CI receiver.

FIG. 3A is a plot showing the steep spectral roll-off of a basic CIpulse.

FIG. 3B shows the spectral response of a Gaussian-shaped pulse.

FIG. 3C illustrates a Gaussian-shaped window applied to a basic CI pulsewaveform.

FIG. 3D is a time-domain plot of a windowed CI pulse.

FIG. 4 illustrates overlapping guard intervals that may be employed in aCI signal.

FIG. 5A illustrates a transmission method and system employingPAPR-reduction signaling of the present invention.

FIG. 5B illustrates a feedback approach for reducing PAPR below apredetermined threshold.

FIG. 6 illustrates a set of methods and components of the presentinvention.

FIG. 7 shows sub-carrier frequencies allocated to individual users ordata symbols.

FIG. 8A illustrates a spreading technique that may be employed in theinvention.

FIG. 8B illustrates basic components of a CI transmitter and a CIreceiver.

FIG. 8C illustrates a receiver system and method of the inventionadapted to process received FD-CI signals.

FIG. 9A illustrates a distribution of time interleaved sub-carrierweights over a plurality of frequency-time channels.

FIG. 9B illustrates a digital-chirp.

FIG. 9C illustrates a distribution of time interleaved sub-carrierweights sub-carrier weights that share the same time slots.

FIG. 10A illustrates a CI transmitter adapted to reduce PAPR ofmulticarrier signals.

FIG. 10B illustrates an embodiment of a CI symbol generator.

FIG. 10C illustrates an alternative embodiment of a CI transmitter.

FIG. 10D illustrates further optional transmitter elements that may beincorporated into various embodiments of the invention.

FIG. 11A illustrates a CI pulse train generated from a superposition ofequally spaced subcarriers.

FIG. 11B illustrates a set of decoding symbols applied to CIsub-carriers.

FIG. 12 is a block diagram of a CI transmission system.

FIG. 13 is a flowchart illustrating the operation of the transmittershown in FIG. 12.

FIG. 14 shows a CI receiver.

PREFERRED EMBODIMENTS

The CI system of the invention provides for control and programmabilityof the frequency spectrum of multicarrier and single-carrier signals. CIsignal synthesis and analysis (i.e., decomposition) may employcombinations of block transforms and sliding transforms, such asdescribed in U.S. patent application Ser. No. 10/414,663, entitled“Orthogonal Superposition Coding for Direct-Sequence Communications,”filed on Apr. 16, 2003, and incorporated by reference herein.

CI provides a common signal-processing platform that can transmit andreceive a broad range of communication waveforms. CI generatesmulti-carrier wavelets used to build communication waveforms and employsorthogonal multi-carrier wavelet bases to process received signals. Eachwavelet is made from simple sinusoids. Accordingly, CI processes allkinds of waveforms as weighted combinations of sinusoids.

One aspect of software-defined CI processing calculates complex-valuedsubcarrier weights in order to synthesize and decompose a variety ofdifferent communication waveforms. In FIG. 1A, a transmitter is providedwith a weight vector (such as any of a plurality of weight vectors 101to 106) selected to generate a predetermined communication waveform. Theweight vector is generated from CI codes that map waveformcharacteristics (e.g., packet length, bandwidth, modulation scheme,etc.) to spectral coefficients. The weight vectors 101 to 106 may begenerated and/or stored in memory. CI coding also spreads a data symbolvector 110 onto the spectral coefficients. The resulting sub-carrierweights are input to the bins of an IDFT or IFFT 115 and thenup-converted to the appropriate RF channel.

A receiver employs a complex conjugate of the appropriate communicationwaveform, as shown in FIG. 1B. A received signal is digitized and downconverted 121 before being processed in a IDFT or IFFT 122. Theresulting spectral components are frequency-domain equalized 123 andvector multiplied by an appropriate waveform-selection vector 111 to 116to extract the coded data symbols. Decoding 124 produces estimates ofthe transmitted data.

In a software-adaptable transceiver module, a CI vector generatorproduces a sub-carrier weight vector corresponding to a particularwaveform. When the vector weights are applied to input bins of aninverse Fourier transform, the resulting weighted spectral components(i.e., sinusoidal waveforms) combine to produce the correspondingwaveform. Different sub-carrier weight vectors can produce waveformscorresponding to different cellular communication standards (such asGSM, IS-95, and UMTS) and different networking protocols (such asBluetooth and 802.11a, b, and g).

In addition to synthesizing and processing conventional waveforms, newsub-carrier weight vectors can be developed to produce future waveforms.This simplifies system upgrades. In order to introduce a new waveform, anew vector can be added. Thus, CI is also adaptable to 3G waveforms,such as MC-CDMA, Spread-OFDM, coherence multiplexing, and UWB.

FIG. 2 illustrates a CI transmitter 200 and a CI receiver 220 that maybe adapted for use in a communications network, such as a mobilecellular network or a wireless local area network. Such communicationsystems consist of a downlink and an uplink. The downlink is typically aunidirectional communication link from a single base station to one ormore remote, and possibly mobile, transceivers. The uplink is typicallya unidirectional communication link from these transceivers to the basestation. Typically, the downlink and uplink employ frequency divisionduplex (FDD) operation in which they occupy distinct non-overlappingfrequency bands.

It is also possible to operate in a time division duplex (TDD) mode inwhich the uplink and downlink occupy the same frequency band butalternate in time. TDD is generally preferred only in systems havingsmall multipath delay spreads. The uplink is a multiple-access channel,since multiple remote transceivers can access and share the uplinkchannel resources. The downlink can be regarded as a broadcast ormulticast link. In general, the problem of interference suppression ismore difficult and also more critical for the uplink, since it typicallyrepresents the capacity bottleneck compared to the downlink.Accordingly, various aspects of the invention are directed towardmitigating interference in CI systems.

The CI transmitter 200 includes a modulation and framing logic 201, a CICoder 202, an optional over-sampling logic 203, an optionalpulse-shaping filter 204, an optional spectrum mask 205, an inverse FastFourier Transform (IFFT) 206, and an transmission module 207. Anoptional framing and overlapping logic (not shown) may coupled to theoutput of the IFFT 206. The CI transmitter 200 may be adapted to accepta spectrum-control instruction to generate the desired spectrum of theoutput signal. In one aspect of the invention, each of a plurality of CItransmitters is synchronized to a reference where the reference isderived from a common source

The input data is sent to the modulation and framing logic 201. Thisblock modulates the binary input data to generate real or complex-valueddata, typically referred to in the art as “constellations”. Themodulated data are framed in blocks of size N. The modulation andframing logic 201 and/or the CI Coder may also include Hermitianextension logic for baseband applications. The Hermitian extension logicadds N/2 conjugate mirror samples to N/2 modulated data samples andforms a data frame of length N. The Hermitian extension is provided forbaseband transmission applications in which the output of the IFFT 206needs to be real-valued.

The modulation of the binary input data can be of many types. Forexample, the modulation can be phase shift keying (PSK), amplitude shiftkeying (ASK), quadrature amplitude modulation (QAM) or other derivativesof these modulation schemes, such as binary phase shift keying (BPSK),quadrature phase shift keying (QPSK) and 64-QAM. The particular type ofmodulation applied to the binary input data is not critical to theinvention.

The CI coder 202 is adapted to encode the data symbols with CI codesthat spread the data over predetermined subcarriers (e.g., selected byfrequency and number of subcarriers). The spreading is typicallyperformed relative to channel conditions on each carrier or on anaggregate of carriers. Each of the subsets of CI sub-carriers may bemodulated with respect to channel conditions of the correspondingsubset. For example, higher modulation constellations may be employedfor subsets characterized by better channel conditions. CI coding mayinclude any combination of coding, including spread-spectrum coding,multiple-access coding, channel coding, and encryption. Other types ofcoding based on CI may be implemented.

After the binary input data has been modulated and framed in frames oflength N, it may be input to the over sampling logic 203. The oversampling logic 203 inserts an integer number M−1 of logic zeros inbetween consecutive data samples. The purpose of the over sampling ofthe modulated data is to increase the frequency resolution of the datain order to allow better control of the spectrum of the CI signalgenerated by the transmitter 200. This forms a newly created frame oflength MN.

If the CI transmitter 200 does not provide for over sampling, thefrequency resolution of the transmitted CI signal is proportional to thegiven bandwidth divided by the frame length N. With over sampling, thefrequency resolution is proportional to the given bandwidth divided bythe frame length MN, which is a factor of M greater. The greaterfrequency resolution allows better precision in controlling the spectrumof the output signal.

The over-sampled data is coupled into an optional pulse-shaping filter204. The pulse-shaping filter 204 performs spectrum shaping of one ormore sub-carriers components of the output CI signal. The pulse-shapingfilter 204 typically shapes the weights that are input to the bins ofthe IFFT 206. Other types of Fourier transforms may be used. Forexample, an IDFT or an I-OFFT may be employed. Other types of invertibletransforms may be employed as appropriate for the output signal form,which depends on how a subcarrier is defined. Accordingly, various typesof sub-carrier generators may be employed in place of the lEFT 206.Various types of subcarriers and corresponding CI processing relating towireless communications are described in U.S. patent application Ser.No. 11/424,176, entitled “Method and Apparatus for Using MulticarrierInterferometry to Enhance Optical Fiber Communications,” filed on Nov.2, 1999, and incorporated by reference herein.

Subcarriers, or subchannels, may include any combination of diversityparameters. For example, each subcarrier may include one or moremulti-frequency components. Each of a set of subcarriers may includeequal and/or different numbers of multi-carrier components (e.g.,carrier frequencies). A set of subcarriers may include a plurality ofcontiguous carrier frequencies and/or non-contiguous carrierfrequencies. A set of subcarriers may include equally spaced and/orunequally spaced carrier frequencies. Each subcarrier may becharacterized by one of a plurality of carrier bandwidths. Similarly,any combination of carrier frequencies and at least one other diversityparameter may be implemented as a subcarrier. Each subcarrier may bedefined by a space-frequency combination.

A spectrum mask 205 may be used to suppress one or more subcarriersand/or sub-carrier sidelobes. The spectrum mask 205 may be adapted toperform sub-carrier allocations to one or more users and/or one or moredata sub-channels assigned to a user.

The pulse-shaping filter 204 may be used to suppress side lobesassociated with predetermined subcarriers of the output CI signal. Thispulse shaping can introduce known patterns of ISI that compromiseorthogonality between the subcarriers. However, a receiver may utilizeequalization, interference cancellation, and/or multi-user detection tomitigate ISI and loss of orthogonality.

A guard-interval module (not shown) may optionally be included in thetransmitter 200. A guard interval unit is adapted to prepend a guardinterval, such as a cyclic prefix, to each subcarrier or one or moregroups of subcarriers. The requirement for cyclic redundancy to mitigatedata corruption caused by multipath may be obviated provided that thelength of a transmitted training sequence (or pilot tone) issufficiently long relative to the channel delay spread. In some cases, atraining sequence may be provided that contains a sufficient number ofchips relative to the channel delay spread. The length of the trainingsequence may be determined by the prevailing dispersion conditionsassociated with the channel.

A guard interval, such as a time-domain redundant cyclic prefix, istypically inserted between transmitted symbols after inverse Fouriertransform processing. In order to simplify frequency-domainequalization, the guard-interval length may be selected such that itexceeds the FIR channel memory. A guard interval may be implemented bypadding trailing zeros (i.e., a null signal) to the end of eachtransmitted symbol, such as described in B. Muquet et. al., “OFDM withTrailing Zeros Versus OFDM with Cyclic Prefix: Links, Comparisons andApplication to the HiperLAN/2 System,” IEEE International Conf onCommunications, New Orleans, LA, Jun. 18-22, 2000, pp. 1049-1053, whichis incorporated herein in its entirety.

Various data-detection algorithms may be employed in a CI receiver if aguard interval is not inserted after every symbol, or if it is omittedcompletely. Such techniques may approximate exact frequency-domainequalization, which eliminates the need for guard intervals. Acomputationally efficient interference cancellation may be performed.Accordingly, C. V. Siun, J. Gotze, M. Haardt: “Efficient Data DetectionAlgorithms in Single—and Multi-Carrier Systems Without the Necessity ofa Guard Period”, ICASSP 2002, Orlando, 2000, pp. III-2737 to III-2740,is incorporated by reference.

Each transmitter may optionally include separate scrambling,interleaving, bit loading, tone ordering, and constellation encoding. Insome applications, specific bit loading or tone ordering can beperformed relative to different information types such that subcarriersare selected for optimal bit rates and error rates.

The transmitter 207 is adapted to process one or more baseband or IFsignals for coupling into a communication channel. The transmitter 207may include various transmission modules and functionality as is wellknown in the art for providing transmission from one or more antennaelements. The output time-domain signal from the IFFT 206 is typicallyconverted into a time-domain analog signal by a conventionaldigital-to-analog converter (DAC). The modulated signals may begenerated and transmitted (and/or received) in a manner to provideantenna, frequency, and/or temporal diversity.

If co-channel interference is localized in frequency, the affected bin-(or bins) having an average signal-to interference-plus-noise ratio(SINR) below a certain threshold can be left unused. If the interferenceis temporary, the bin can be reused when the SNR improves. Thisprocedure is well known in digital subscriber line modems that use DMTas the modulation scheme. This procedure may be implemented in awireless multi-carrier system by measuring co-channel interferenceacross the frequency band of interest. Furthermore, fading causesfluctuations in the channel frequency response with respect to time.Thus, carriers experiencing deep fades may be avoided.

Allocation of frequency bins (i.e., determining the spectral locationsor bin numbers) to different users may be performed by taking spatialand other information (such as automatic gain control (AGC) information)into account. Each user may require one or more bins to meet a certainquality of service requirement. This aspect of the invention isparticularly appropriate to adaptive-array architectures. However, it isnot limited to such receiver configurations, as will be apparent tothose skilled in the art.

The bins allocated to any one user (or any one data stream correspondingto a subset of a user's subcarriers) may be spaced far apart infrequency to minimize mutual inter-bin interference and enhancefrequency diversity. The allocation of frequency bins to users and datasub-channels in multi-antenna systems is of particular importance. Eachbin may be placed in a spectral location with bins belonging to otherusers that are spaced as far apart as possible in the dominantdirections of arrival (DOAs) of their signals. Each bin may be placed ina spectral location with bins belonging to other users such thatdifferences in signal strength of active bins in the neighborhood areminimized.

Each bin may be placed in a spectral location such that there are noco-channel interferers in the same frequency band. If no such locationsare available, spectral locations can be chosen based on the DOA andsignal strength of co-channel interference. In general, co-channelinterference bins with lower signal strength are assigned before binswith higher interference signal strength. Also, bins are preferablyallocated so that the difference in the DOA's of the particular user andthe interference are as large as possible. These criteria are balanceddepending upon the operating environment. These criteria may be weightedto construct algorithms or flowcharts optimized for specific channelconditions and interference scenarios, as will be apparent to thoseskilled in the art.

Sub-channel allocation and control may be implemented with a statustable for each bin. The table may include information, such as whethereach bin is active or inactive at a given time, which user is assignedto the bin, type of data, modulation scheme and constellation size ofsymbols in the bin, received power level, dominant DOA's of the useroccupying the bin, power level and DOA's of any co-channel interferersspectrally overlapping with the bin, etc. Note that all items of theabove information may not be available or be able to be computed foreach bin at all times. Some of the status items (such as received powerlevels) may be updated periodically. Other items may be updated via anevent-driven mode, such as user activity and network control functions.

CI-based signaling can provide high spectral efficiency to all types oftransmission protocols. In the frequency domain, the subcarriers areoften overlapping and orthogonal. In conventional single-carrierprotocols implemented with CI, the spectral roll-off at the band edgesis shaped with respect to the long symbol period of the CI symbols,which is typically much longer than the symbol period of a conventionalsingle-carrier signal. Therefore, the roll off is much steeper in CIsignaling than it is for conventional single-carrier signaling. Unlikeall other multi-carrier protocols, CI provides a low PAPR because thesuperposition of CI sub-carriers resembles a single-carrier signal.

FIG. 3A illustrates a nearly rectangular spectral response 301 of a CIpulse 311 (shown in FIG. 3D) having ten subcarriers (modulated withrandom data) compared to the spectral response 302 of a Gaussian-shapedpulse 312 (shown in FIG. 3B) having the same pulse width as the CI pulse311. The CI pulse 311 and the Gaussian pulse 312 are characterized by afrequency offset (e.g., an IF or carrier frequency) f₀ of 20, whereinthe units of frequency are shown in normalized, generic form (and may beshifted to a desired frequency band).

The relationship between the CI and Gaussian waveforms in both the timedomain and the frequency domain characterize trade offs that can be madebetween PAPR and data-bearing capability (i.e., BER, spectralefficiency, etc.). The CI pulse 311 has higher sidelobes in the timedomain than the Gaussian pulse, which contributes to higher PAPR. The CIspectral profile 301 has a steeper roll off (and thus, better spectralefficiency) compared to the Gaussian spectral profile 302.

FIG. 3C illustrates a Gaussian-shaped window 303 applied to a basic CIpulse waveform 308 over a symbol duration T_(s) that produces the CIpulse 311 shown in FIG. 3D. In contrast, the Gaussian pulse 312 has asymbol duration of T_(s)/N, where N=10. The CI pulse waveform 308 has aspectrum consisting of 10 equally spaced narrowband subcarriers. Thewindow 303 is characterized by a spectral bandwidth that is inverselyproportional to its duration. If the subcarriers are contiguously (andequally) spaced with respect to the inverse of the symbol durationT_(s), The window spectrum convolved with the CI pulse spectrum reducesthe roll off of spectral response 301 while flattening the CI pulse 311sidelobes. Similarly, frequency-domain shaping of the CI sub-carrierspectrum may be performed, as is well known in the art. Alternativeembodiments of the invention may employ different window shapes thanshown in the drawings and described herein.

Various arrangements of CI sub-carriers are possible, such asnon-contiguous and unequally spaced sub-carrier sets. Equivalently,different symbol periods T_(s) relative to sub-carrier spacing(s) may beprovided. In some aspects of the invention, different subcarriers and/orsub-carrier sets may be provided with different symbol durations. Thespectrum of the resulting signal will vary accordingly. However, thespectral-efficiency advantages of CI are typically preserved. Forexample, when subcarriers of a particular channel or subchannel areinterleaved with other subcarriers, the sub-carrier spacing may bearranged such that multiple channels or subchannels are orthogonal toeach other, even if their sub-carrier spectra overlap.

It is well known in the art to apply windowing to smooth the transitionsbetween adjacent data symbols, and thus, increase the spectral roll off.The following cyclic prefix shaping technique is well known anddescribed in Chapter 2 of R. Van Nee and R. Prasad (OFDM for WirelessMultimedia Communications, Norwood, Mass. Artech House, 2000, pp.33-52). A cyclic prefix and postfix are added to each symbol. Theresulting extended symbol is typically windowed using a Nyquist window(or some other common window) wherein the roll off is selected such thatall of the prefix and postfix are windowed, but the original symbol isleft unchanged. Adjacent symbols are then overlapped so that thewindowed cyclic prefix of a particular symbol is added to the windowedcyclic postfix of the preceding symbol.

In F. Giannetti, “OFDM Communications Primer”, Intellon Corporation.White Paper #0032, March 1999, which is incorporated by reference, thecyclic prefix and postfix are shaped using a raised-cosine window toprovide smooth transitions at the band edges in the time domain, whichincreases the slope of the spectral band edges.

FIG. 4 illustrates overlapping guard intervals (such as shown in J. M.Paez-Borrallo, “Multicarrier vs. Monocarrier Modulation Techniques: AnIntroduction to OFDM”, BWRC Retreat—Winter 2000, Berkeley WirelessResearch Center, Berkeley, Calif. Jan. 9-11, 2000, which is incorporatedby reference) that may be applied to CI signals. In this case, an extraguard time XGI includes an overlap of a postfix and an extended cyclicprefix CP. Signals in the extra guard time are shaped to smoothtransitions between each symbol and the following cyclic prefix. Thishelps to minimize adjacent channel interference.

A guard interval may be added to a superposition of CI subcarriers.Alternatively, a guard interval may be provided to each of a pluralityof superposition signals produced from subsets of CI subcarrierscorresponding to one or more users. In some cases, a guard interval orcyclic prefix may be generated and applied to each individualsubcarrier, as is well known in the art and described in A. Czylwik,“Comparison between adaptive OFDM and single carrier modulation withfrequency domain equalization”, IEEE Vehicular Technology Conference,pp. 865-869, Phoenix, AZ, May 1997, and J. Tubbax, et. al., “OFDM versusSingle Carrier with Cyclic Prefix: a system-based comparison,” Proc. ofIEEE Vehicular Technology Conference, Vol 2, pp. 1115-1119, Oct. 2001,which are hereby incorporated by reference.

The present invention may employ Offset Quadrature Phase Shift Key(OQPSK) modulation, which is well known in the art. Various types ofmodulation may be applied to either or both the individual subcarriersand one or more sub-carrier superpositions. In OQPSK, an odd bit streamis delayed (typically by half a bit) with respect to an even bit stream.Minimum Shift Key (MSK) is derived from OQPSK by providing a tapered(e.g., a half-cycle sinusoidal) pulse. An MSK signal is defined as:S(t)=d(t)[W _(I)(t)cos(2πft)+W _(Q)(t)sin(2πft)]where W_(I)(t) and W_(Q)(t) are in-phase and quadrature-phase windows.MSK modulation provides a linear phase change and limited to apredetermined phase shift. Various forms of continuous phase modulationmay be employed in the invention. In Gaussian Minimum Shift Key (GMSK)modulation, a Gaussian-shaped impulse response filter generates thewindow functions W_(I)(t) and W_(Q)(t). GMSK may be employed via FSK orQPSK modulation. Combinations of FSK and QPSK may also be used.

Modulations used in the present invention may include variations andcombinations of modulation schemes mentioned in the followingpublications, which are incorporated by reference in their entirety:“Electromagnetic compatibility aspects of radio-based mobiletelecommunication systems (DTI Link report, 1999)”, Produced in 1999 forthe LINK Personal Communications Programme by ERA Technology Ltd,Leatherhead, Surrey, KT22 7SA, England, “Practical GMSK DataTransmission”, MX Com, Inc., Application note 1998, reprinted fromWireless Design and Development, January 1995, Vol. 3, No. 1, H. Zou et.al., “Equalized GMSK, Equalized QPSK and OFDM, a Comparative Study forHigh-Speed Wireless Indoor Data Communications”, J. Vankka, “Amulticarrier GMSK Modulator”, IEEE J. Sel. Areas in Comm., Vol. 19, No.6, June 2001, and J. Vankka et. al., “A Multicarrier GMSK Modulator forBase Station”, ISSCC 2001, Session 22, Technology Directions: System ona Chip/22.7.

In one aspect of the invention, MSK modulation is employed forsub-carrier symbols and/or data symbols modulated onto sub-carriersuperpositions. The symbols preferably have symbol durations T_(d)greater than the duration T_(s) required for orthogonality so as toprovide a guard interval. Continuous frequency variation of aguard-interval waveform is provided as needed within each guard intervalto match the phase of each corresponding adjacent symbol waveform.Continuous amplitude variation is provided to a guard-interval waveformas needed within each guard interval to provide a smooth transitionbetween the amplitudes of the corresponding adjacent symbol waveforms.In the case wherein digital implementations of the symbol andguard-interval waveforms are provided, the guard-interval waveforms areconstructed to approximate either or both the continuous frequencyvariation and the continuous amplitude variation.

Each channel data stream, each sub-channel, each antenna, or some otherunit of transmission can be modulated with a particular modulationscheme, such as selected from a set that includes M-PSK and M-QAM.Channel coding can be provided to each channel data stream, eachsub-channel, each antenna, etc. Pre-conditioning of the data may also beperformed at the transmitter unit using channel state information thatis descriptive of the characteristics of the communications links. Suchstate information may include, for example, carrier to interferenceratios or eigenmodes corresponding to the communications links

Each subcarrier is phase and/or amplitude modulated, typically via PhaseShift Key (PSK) or Quadrature Amplitude Modulation (QAM), in which eachsymbol value is represented by a point in the complex plane. The numberof available symbol values depends on the number of bits in each symbol.During initialization of a communications session, the number of bitsper symbol (i.e., bit loading) for each subcarrier or predeterminedgroup of subcarriers may be determined according to the quality of thecorresponding transmission channel. In the case where CI-coded sets ofCI sub-carriers are generated, bit loading on each sub-carrier setcorresponds to the channel quality corresponding to the set. Thus,different sets may be provided with different data-symbolconstellations. Similarly, sub-carrier allocations for each set maydepend on given channel conditions, as well as other considerations,such as data rate. Sub-carrier allocations may vary with respect tosub-carrier sets. For example, sub-carrier sets may have differentnumbers of subcarriers, different sub-carrier frequency spacings,different sub-space profiles (e.g., transmitted by different groups ofantennas), as well as other sub-carrier signal characteristics.

The channel quality is typically affected by frequency-dependentattenuation (e.g., multipath fading, scattering, absorption, path loss,etc.) and/or interference (e.g., co-channel interference, inter-symbolinterference, multiple-access interference, etc.). Various types ofsignal distortion may occur in non-linear channels. Channel quality istypically measured as SNR, SNIR, BER, Packet Error Rate, Probability oferror, etc. High-quality subchannels may be used to communicate datawith a relatively dense QAM constellation, whereas low-quality channelsmay be limited to smaller constellations that allow a greater distancebetween adjacent points in the constellation. Some allocated subchannelsmay not be loaded with any bits due to attenuation and/or interferencein those channels.

Multi-carrier modulation permits much of the data processing to becarried out digitally. Typically, the incoming bit stream is seriallyreceived and then arranged into data symbols. Reed-Solomon coding andother coding techniques are typically applied for error detection andcorrection. In CI, the coded data symbols are provided with polyphasecoding that spreads each data symbol over multiple CI code symbols. EachCI code symbol is applied to each subcarrier.

In one embodiment of the invention, the data symbols are modulated ontoa plurality of sub-carrier sets. Subcarriers in each sub-carrier set areprovided with CI coding to minimize the PAPR of the composite, orsuperposition, signal. Other forms of PAPR reduction may be employed.For example, the unloaded subchannels may be used to transmitPAPR-reduction symbols. Similarly, unallocated and/or unused channelsmay be provided with PAPR-reduction symbols. The composite signals areamplified via power amplification prior to being combined.

Modulation of the subcarriers can be performed by applying a Fouriertransform operation, such as an inverse Discrete Fourier Transform(IDFT), to the encoded symbols, producing an output modulatedtime-domain signal. The modulated signal is then serially transmitted.Much of the processing operations may be implemented on a single chip,such as a DSP, FPGA, or an ASIC.

The analog signals produced by the transmitter 200 shown in FIG. 2 arecommunicated over the channel to the receiver 220, which reverses thetransmission process to recover the transmitted data. A communicationschannel may include any type of medium or mechanism for providing datafrom a transmitter to a receiver. The non-ideal impulse response of thetransmission channel distorts the transmitted signal. Accordingly, thesignal received by the receiving modem is a convolution of the analogtransmit waveform with the impulse response of the channel. Ideally, thereceived subchannels are orthogonal such that the modulated data can beretrieved via Fourier transform (e.g., a DFT or FFT) demodulation orsome other invertible transform characterized by the appropriateorthogonal basis functions.

Signals coupled from the channel are processed in a receiver module 227.The signals are optionally filtered by an analog filter (not shown) inthe receiver module 227. Filtering may be provided for anti-aliasingand/or to remove high-frequency noise from the received signal.Equalization of the analog signal may also be performed to compensatefor attenuation in the transmission channel. An analog-to-digitalconversion module (not shown) may be provided to convert the filteredanalog signal into the digital domain. A digital filtering module (notshown) may be employed to augment the function of the analog filter (notshown).

A time domain equalizer (TEQ) 226 may optionally be provided. The TEQ226 is preferably a finite impulse response (FIR) digital filter, whichmay be implemented as a software routine executed by a DSP. The TEQ 226can be designed to effectively shorten the length of the impulseresponse of the transmission channel, including the filtering that isperformed prior to receipt by TEQ 226. The design of a TEQ filter mayinclude selecting FIR coefficients during initialization or trainingoperations associated with establishing a communications session.

A guard-interval removal module (not shown) may precede an FFT 225. Thecomplex frequency values produced by the FFT 225 are processed by afrequency domain equalizer FEQ 224. The FEQ 224 scales the values withan appropriate vector of complex numbers that ensures phase and powerequalization of the received signal. The FEQ 224 vector may be adapteddynamically and/or set during an initialization process to compensatefor phase distortion and power attenuation introduced by the channel.The equalized complex values are despread into data symbols by a CIdecoder 223. Channel decoding (e.g., Reed-Solomon decoding andde-scrambling) in decoder 222 completes the receiver data processing.The resulting bits are streamed as appropriate for the application.

The baseband processing portion of the receiver may include apilot-tracking loop (not shown). The loop may incorporate a phase errormetric utilizing a maximum likelihood estimator for phase errors of thereceived data symbols.

Channel compensation, such as performed by the TEQ 226 and the FEQ 224,depends on an effective channel estimation algorithm. Channel estimationtypically includes any technique used to characterize a time domainand/or frequency domain channel response. Pilot-based channel estimationtypically tracks the amplitude and phase of at least one transmittedpilot tone. Similarly, training symbols may be transmitted preceding adata payload, following a payload, or interleaved into a payload.

Interpolation may be provided between adjacent channel estimates totrack changing channel conditions. In some aspects of the invention,training symbols are placed in the header and trailer of each packet.Training symbols may even be placed in the data payload. Channelestimates are then interpolated over the interval between the trainingsymbols. For example, channel estimates over a packet duration may beestimated by assuming a linear variation in the channel between two ormore channel estimates. Estimated channel characteristics may begenerated in intervals containing no training symbols by providing alinear interpolation function between the measured training symbols.Interpolation functions may be developed to more closely characterize achannel's rate of change. Other interpolation functions may be used toestimate channel variations.

Alternatively, blind or semi-blind algorithms may be employed forchannel estimation. Accurate channel estimation is possible through theuse of Wiener frequency domain MMSE deconvolution combined withfrequency domain spatial decoupling matrices.

In a multi-carrier system, channel estimation typically involvesdetermining channel impulse responses of a plurality of sub-carrierchannels. For example, transform operations may be performed on both atraining sequence received from the communications channel and a replicaof the received sequence that is generated locally. In this case, thetransform operations are arranged to generate a plurality of frequencybins corresponding to the received training signal and a plurality offrequency bins corresponding to the locally generated signal. Thetransform operations are followed by point-by-point operations betweencorresponding received signal frequency bins and local signal frequencybins. Concatenating the point-by-point operations associated with thechannel provides a composite frequency response for the channel. Thiscomposite frequency response allows the generation of a time domainchannel impulse response for the channel. Other types and variations ofchannel estimation may be performed.

The channel estimator may utilize the outputs of the FFT 225 todetermine an initial course frequency estimate from short symbols of apreamble and a fine frequency estimate from long symbols of thepreamble. This information is used to generate the coarse and finefrequency estimate signal, such as employed in a pilot-tracking loop.Alternatively, pilot tracking may occur before the FFT 225 operationsuch that the phase error for subsequent data symbols is reduced priorto processing by the FFT 225.

The step of equalizing includes a step selected from the groupconsisting of: performing a frequency domain equalization, performing atime-domain equalization, performing a decision feedback equalization,performing an iterative equalization, performing an ISI cancellation,performing a turbo equalization, and performing a maximum likelihoodsequence estimation. Fractionally spaced equalization may be adapted tosystems employing cyclic prefixes.

Coefficients for use in an equalizer filter are typically based on thenoise covariance and channel impulse response of the correspondingcommunications channel. A channel impulse response is determined fromthe channel transfer function, and a noise covariance is based on thenoise power spectral density. These channel characteristics aretypically measured from known transmissions or calculated from receiveddata.

Sub-space algorithms, such as semi-blind subspace-based channelestimation, have been shown to reduce variations in channel tracking.Equalization may be based on any known equalization scheme, such asoverlap-and-add block convolution or overlap-and-save algorithms.

The communications receiver includes at least one of a time-domainequalizer and a frequency-domain equalizer. The receiver may furtherinclude an update mechanism configured to update the time-domainequalizer and/or the frequency-domain equalizer based upon measured orestimated performance of a communications link from which the receiverreceives data.

An equalizer typically uses an algorithm with a set of coefficients thatare selected based on noise power and an impulse response of thecommunications channel. An estimate of the original data is recovered bydemodulating the equalized data. The coefficients are selected accordingto one or more particular approaches. For example, the coefficients maybe selected to optimize the impulse response length of thecommunications channel to reduce interference, such as inter-symbolinterference (ISI) and inter-channel interference (ICI), to reduce noisepower, or to simultaneously optimize impulse response length and reducenoise power. As another example, the coefficients may be selected toreduce noise power due to ISI, ICI, and noise from additionalinterference sources, including, but not limited to, inter-cellinterference, jamming, and/or other external sources of interference. Asyet another example, the coefficients may be selected by minimizing afunction of a channel's impulse response and noise power, or noise powerspectral density.

An equalization technique may employ a Wiener least squares filter thatutilizes a modified inverse filter to control the white noise responseof the filter, i.e. the undesired enhancement of thermal noise from theantenna. In M. Haardt, “Smart Antennas for Third Generation Mobile RadioSystems”, Sixth Annual Workshop on Smart Antennas in WirelessCommunications, Stanford University, Palo Alto. Calif., July 1999, whichis incorporated by reference, channel equalization is described in termsof a Wiener filtering response. Channel equalization in the context ofthe Wiener filter is presented in H. Sari et al., “TransmissionTechniques for Digital TV Broadcasting”, IEEE Communications Magazine,Vol. 33, No. 2. pp. 100-109, February 1995, which is incorporated byreference.

Various types of multi-user detection may be employed in the presentinvention. These types include hard decision and soft decisionmulti-user detection techniques. Multi-user detection techniquesdescribed in the following publications are incorporated by reference:J. Blantz et. al., “Performance of a cellular hybrid C/TDMA mobile radiosystem applying joint detection and coherent receiver antennadiversity”, IEEE J. Selected Areas in Comms. 12(4), May 1994, pp.568-579and A. Klein et al., “Zero forcing and minimum mean square errorequalization for multiuser detection in code-division multiple accesschannels”, IEEE Trans. Veh. Tech. 45(2), May 1996, pp.276-287.

The length of the impulse response of the physical channel determinesthe required length of the guard interval. However, a long guardinterval reduces the effective throughput of the transceiver. Thus, toavoid using a long guard interval, filters may be employed to shortenthe channel impulse response and thereby allow the use of a shorterguard interval. Time domain linear filters, often referred to asshortening impulse response filters (SIRFs) or time-domain equalizers,are typically employed to shorten the channel impulse response.

A number of techniques have been proposed for designing TEQ filters. J.W. P. Melsa and R. C. Younce, “Impulse Response Shortening for DiscreteMultitone Transceivers,” IEEE Trans., COM-44, (12), 1662-1672 (1996);and N. Al-Dahir and J. M. Cioffi, “Stable Pole-Zero Modeling of Long FIRFilters With Application to the MMSE-DFE,” IEEE Trans., COM-45, (5)508-513 (1997), are incorporated by reference herein. Generally, thesefilter design algorithms are typically based on least mean square (LMS)or eigenvector calculus.

Filter coefficients for SIRF filters may be selected in order to satisfyconstraints in both the time and frequency domains to improveperformance. More specifically, SIRF filters may be adapted to shortenthe channel impulse response in the time domain while also providing afrequency response that does not attenuate or amplify the receivedsignal.

An SIRF filter satisfying both frequency and time constraints isobtained by determining the intersection of constraint sets that must besatisfied in both time and frequency domains. By varying the setsutilized to define the time and frequency domain constraints, SIRFfilters having a linear or non-linear phase response may be obtained. Avector space projection method or equivalent iterative algorithm may beapplied to the sets until the algorithm converges to a solution (i.e.,either the intersection or a point with minimum summing distance to allsets). In the case of SIRF design, the solution of the algorithmprovides the filter coefficients.

In one aspect of the invention, a function that may be performed duringthe training sequence on initiation of a communication sessiondetermines the number of bits per symbol (i.e., the bit loading)assigned to each subchannel. After bit loading, it is common for anumber of subchannels (typically sub-channel frequencies experiencingdeep fades or interference) to remain unloaded, carrying no data symbolsat all. Similarly, there may be other unused channels, such asunallocated channels. The PAPR is reduced by using these unloadedsubchannels to carry a PAPR-reduction signal that does not necessarilycontain any data payload, but has the effect of reducing the amplitudeof the time-domain signal below a predetermined PAPR threshold. Channelsreserved for other purposes may be used for PAPR-reduction signaling.Channels may be allocated specifically to convey PAPR-reduction signals.

In one aspect of the invention, a PAPR-reduction circuit coupled to acarrier generation circuit in a transmitter is adapted to produce one ormore PAPR-reduction signals. The PAPR-reduction signals are combinedwith the data-bearing subcarriers prior to amplification by one or moreamplifier circuits. The amplified PAPR-reduction signals may optionallybe filtered, or otherwise attenuated, prior to coupling into acommunication channel. An optional attenuation module may be adapted toprovide frequency-dependent attenuation to the amplified signals, suchas to reduce or remove one or more amplified PAPR-reduction signals. Thegenerated PAPR-reduction signals may be used to ensure low PAPR ateither or both the transmit side and the receive side.

In another aspect of the invention, a PAPR-reduction circuit is coupledto a low-noise amplifier in a receiver. The PAPR-reduction circuit isadapted to lower the PAPR of a combined signal consisting of at leastone received signal and at least one PAPR-reduction signal. Thereceiver's PAPR-reduction circuit may be adapted to generate one or morePAPR-reduction signals from any combination of one or more receivedPAPR-reduction signals transmitted by a remote transceiver, one or morepilot tones, one or more training signals, one or more control signals,and one or more data-bearing signals.

In another aspect of the invention, methods are provided that perform aniterative process to derive symbols for one or more unloadedsubchannels. For example, an initial trial value (possibly zero) of theunloaded sub-channel signal is added to the time-domain signal generatedby the IDFT. A non-linear function corresponding to the power amplifieris applied to the combined time-domain signal. If no clipping occurs(i.e., if none of the signal elements change), the current trial valueof the unloaded sub-channel signal is kept as part of the signal.Alternatively, if the nonlinear amplifier function indicates clipping,the clipping is used to determine a new trial signal for the unloadedsubchannels, and the process is repeated until clipping does not occur.

FIG. 5A illustrates a transmission method and system employingPAPR-reduction signaling of the present invention. A transmitting modemis provided with an input bit stream that is to be transmitted to atleast one receiving modem over a wireless communication channel. The bitstream is coupled to a symbol-mapping module 501. The mapping module 501groups the sets of input bits to multiple-bit symbols. The number ofbits allocated to each subchannel may vary with respect to thecorresponding bit loading determined in an initialization step. Thesymbols correspond to points in at least one modulation constellation(e.g., QAM, PSK, etc.).

The mapping module 501 may optionally include a channel coder 511 (e.g.,an error-correction coder) adapted to provide coding for error detectionand correction. The channel coder 511 may be adapted to perform anycombination of various types of coding, such as Reed-Solomon coding,trellis coding, turbo coding, CI coding, and/or LDPC coding. The mappingmodule 501 preferably includes a CI coder 521 adapted to perform atleast one predetermined combination of spreading and channel coding. Thesymbols generated by the mapping module 501 are typically complexsymbols including both amplitude and phase information. The mappingmodule 501 is typically adapted to provide a serial-to-paralleloperation on the symbols.

The encoded symbols are then applied to a carrier-generator module 502(such as a system or module adapted to perform an IDFT or IFFT). Themapping module 501 may optionally apply a plurality of differentsequences of a given data block to different subcarriers as part of aprocess for identifying a sequence that minimizes clipping or PAPR. Thecarrier-generator module 502 associates each input symbol with one of apredetermined set of subchannels, and generates a correspondingtime-domain sequence. The carrier-generator module 502 is typicallyadapted to perform a parallel-to-serial operation for generating thetime-domain sequence.

The carrier-generator module 502 converts the input digital bit streaminto a serial sequence of symbol values corresponding to a superpositionof the predetermined set of modulated sub-channel frequencies. Themodulation is indicative of the various data values. Typically, N/2unique complex symbols (and its N/2 conjugate symmetric symbols) in thefrequency domain will be transformed by an IDFR function into a block ofN real-valued time domain samples.

An optional guard-interval module 503 adds a guard interval (e.g., acyclic prefix or a null signal) to each block of serial samples. A guardinterval has the effect of limiting inter-symbol interference (ISI) dueto energy from a previous symbol spreading into the next symbol due tothe channel response. A cyclic prefix causes the data stream to appearto be periodic over a block of N of the N+P samples (where P is thelength of the prefix) such that the equivalence between frequency-domainmultiplication and time-domain convolution is valid.

An optional up-sampling module 504 and an optional digital filter 505may process the digital data stream. Up sampling inserts zero-valuedsamples between each of the input signal samples to increase the samplerate by some predetermined multiple. The digital filter 505 may includea digital low-pass filter to remove image components and/or a digitalhigh-pass filter to remove interference. The digitally filtered signalis then processed by a DAC 506, which produces an analog signal. Ananalog filter (not shown) may optionally filter the output analogsignal. The analog signal is amplified by a power amplifier 507, whichoptionally performs a clipping function, such as a hard-limitingclipping function that limits the signal amplitude. The DAC 506,amplifier 507, and any analog filter may be implemented in acoder/decoder (codec) integrated circuit.

The amplified analog output is then coupled into a transmission channelby a channel coupler 508, such as one or more antenna elements. Thecoupler 508 may include transmit circuitry, as is typically provided inRF transmitters. In general, the receiving modem (not shown) reversesthe processes performed by the transmitting modem to recover the inputbit stream as the transmitted communication.

An unloaded channel-encoding module 512 generates symbols that areassigned to unloaded subchannels in order to reduce the likelihood ofclipping. The symbols are generated and applied to reduce the PAPR ofthe transmitted signal. A clip-prevention signal can be generated thatis orthogonal to the data-payload signal by allocating theclip-prevention signal to subchannels where the data-payload signal iszero. To reduce the peak amplitude of the data-payload signal aftercarrier generation (IDFT) and D/A conversion, the clip-prevention signalis adapted to maintain the data-payload signal below the clippingthreshold. The clip-prevention signal may be applied before or aftercarrier generation.

A particular benefit of the methods of the invention is that thereceiver need not transform the signal, and that little or no data rateis lost. The modifying signal applied by the transmitter only affectsunloaded (e.g., deeply faded) subchannels, and is not detected orconsidered by the receiver when demodulating the transmitted signals onhigher-quality subchannels.

When low-quality channels are sacrificed to provide for transmission ofa clip-prevention signal, the total transmit power can be increased,thereby increasing the throughput (e.g., by enabling larger modulationconstellations and/or reducing channel coding). The increased throughputtypically more than offsets the loss due to sacrificing the low-qualitychannels.

In cases in which CI signaling is provided by the transmitter, unloadedsubchannels may be combined to generate superposition signals for PAPRreduction, or clip prevention. The superposition signals may be used toreduce the amplitudes of constructive superpositions ofinformation-bearing CI pulses. For example, the superposition signalsmay be used to reduce peaks resulting from superpositions of CI pulsesidelobes with other side lobes and/or CI pulse main lobes. Similarly,CI signaling may be provided on the unloaded subchannels, even whenother types of multi-carrier modulation are used. The Cl-pulse waveformsprovide for a low-complexity solution to canceling peaks generated byother multicarrier transmission protocols.

The present invention may be implemented via an iterative approachexecuted in a transmitter feedback loop, as illustrated in FIG. 5B. Atraining process 531 may optionally involve channel estimation toidentify subcarriers characterized by poor channel quality (e.g., deepfades, interference, distortion, etc.). Channels that are unsuitable fordata communications (or minimally suitable with respect to somepredetermined quality-measurement threshold) may be allocated fortransmission of one or more PAPR-reduction (e.g., clip-prevention)signals.

In a MIMO configuration, subcarriers may be identified with respect tounique space-frequency combinations. Thus, certain sub-space channelsidentified as unsuitable for data communications may be allocated forPAPR-reduction transmissions. Similarly, unallocated subchannels and/orallocated (but unused) subchannels may be identified for PAPR-reductiontransmissions. In some cases, subchannels in a network (such as a MIMO,MISO, or SIMO network) may be allocated specifically for PAPR-reductiontransmissions.

An initial trial value of an orthogonal clip prevention signal isselected in a clip-prevention initiation step 532. The initial value maybe zero. The initial value may be retrieved from a look-up table, or itmay be derived from an amplitude-profile estimation of the unalteredtransmission signal. The clip-prevention signal is orthogonal to thedata in that it is applied to orthogonal subchannels or subspaces.

In step 533, a time-domain signal is produced by performing an IDFT ofthe payload data signal and the trial value of the clip-preventionsignal. A nonlinearity corresponding to the clipping threshold may beapplied to the resulting superposition signal. The time-domain signalmay be generated via CI processes.

In step 534, the resulting time-domain signal is examined to determineif any clipping has occurred (or will occur). If clipping occurs, a newestimate for the clip-prevention signal is generated 535. The newnonlinear clipping function is applied to the data. In some cases, theclip-prevention signal is applied to the IDFT with the data. Thetime-domain clip prevention signal may be combined with the time-domainsignal generated from an IDFT of the data. If there is no clipping, themost recent trial value of the clip-prevention signal is maintained andis transmitted in the unloaded channels along with the DMT-modulatedpayload 536.

Different embodiments of the invention may be provided for mapping eachof a plurality of data symbols to a plurality of subcarriers. Spreadingmay include any combination of signal-processing techniques that map adata symbol into a plurality of symbols, such as spread-spectrum coding,channel coding, redundancy, etc. Channel coding typically includes anyof various combinations of block coding, convolutional coding, andparity check coding. Implementations of channel coding may provide foriterative processing, feedback techniques, and various types of decisionprocessing, including hard and soft decision algorithms.

Although subcarriers shown herein are basic multi-frequency subcarriers,different types of channelization (including combinations ofchannelization techniques) may be employed. For example, subcarriers maybe interleaved in time or frequency hopped. The subcarriers may employspace-time channels, space-frequency channels, code-space channels,and/or any other forms of sub-space channelization. Various codingtechniques, as well as other signal-processing techniques, may beemployed to provide for sub-space processing and related functionality,such as successive interference cancellation.

The inventive aspects (i.e., the functionality) of the methods and thesystem components illustrated and described throughout the specificationmay be provided by different combinations of steps and components (andin some cases, by entirely different steps and/or components). Theillustrated and described embodiments herein are meant to merely conveythe basic functionality of the invention.

FIG. 6 illustrates a set of methods and components of the presentinvention. A data input port 601 is provided for accepting a plurality Kof data bits 611. A coding module 602 spreads the data bits 611 toproduce a plurality of coded data bits 612 with M-fold spreading (i.e.,each data bit d_(k) is mapped to M coded data bits

_(m). An interleaving module 603 and a mapping module 604 group thecoded data bits into a plurality N′P of data-bit sets and maps each setinto a plurality N′P of data symbols 614. A second coding module 605spreads the data symbols into a second set of coded data symbols 615.

Each row of symbols shown in set 615 may be converted into one or morenew coded data symbols. For example, a new coded data symbolcorresponding to each row in set 615 may include a complex weightapplied to a subcarrier. In another aspect of the invention, symbols ineach row are mapped to a plurality of subcarriers via some combinationof spreading and/or channel coding.

The coded data symbols are provided to a sub-carrier generator, such asan IDFT 606. The IDFT 606 shown in FIG. 6 is adapted to generate aplurality of communication waveforms, wherein each waveform ischaracterized by a different set of sub-carrier frequencies 616. Symbolsin each row in set 615 are mapped (e.g., spread) onto the subcarriers ofthe waveform corresponding to that row.

Additional signal processing that is applicable to single carrier ormulticarrier signaling may be performed. For example, a guard-intervalmodule 607 may be provided to add a cyclic prefix to the individualsubcarriers, to sub-carrier groups, and/or to the transmissionconsisting of a combined set of all of the subcarriers. A transmitmodule 608 is adapted to provide all digital and analog processingnecessary to adapt the multicarrier signal for transmission into acommunication channel. The transmit module 608 may provide poweramplification, filtering, up conversion, DAC operations, as well as anyother transmitter-related functions to individual subcarriers and/orgroups of subcarriers.

FIG. 7 illustrates an exemplary spectral distribution of subcarriersemployed in the invention. A first set of subcarriers 731.1 to 733.1 areallocated to a particular user. A number P−1 of additional sub-carriersets, such as set 731.P to 733.P, are allocated to the same user. In oneparticular aspect of the invention, subcarriers allocated to aparticular user are divided into a plurality P of sub-carrier sets (suchas illustrated in FIG. 7) in order to convey symbols on differentfrequency division channels. This channelization technique can greatlysimplify some of the transceiver operations, such as coding, decoding,multi-user detection, etc.

Improved frequency-division techniques employed in multi-carrier systemsare described in B. Natarajan et. al., “Introducing Novel FDD and FDM inMC-CDMA to Enhance Performance”, Proceedings of IEEE Radio and WirelessConference 2000 RAWCON'00), pp. 29-32, Denver, Colo., Sep. 10-13, 2000and PCT Appl. No. PCT/US99/02838, which are incorporated by reference.

Subcarriers allocated to a particular user may include contiguous ornon-contiguous sets of subcarriers. Each set of subcarriers preferablyspans a wide frequency band such that if there are N′ subcarriers ineach set, there can be up to N′-fold frequency diversity. However, sincethe number N′ of subcarriers in each set is a fraction of the totalnumber of subcarriers N=N′P allocated to a user, a loss of a subcarrier(e.g., due to a deep fade or interference) substantially distorts thesignals in the corresponding set. Loss of several subcarriers in a setcan result complete loss of the data in that set. Thus, the spreadingaspect of the invention, such as illustrated in FIG. 6, can provide theperformance benefits of spreading each data bit over a large number ofsubcarriers to the reduced complexity of processing small sub-carriersets.

In FIG. 6, N=N′P sub-carriers are allocated to a particular user inwhich there are P coded sub-carrier sets of N′ subcarriers each. Themapping module 604 maps L bits into each symbol to generate N′P symbolsper map set. The coder 605 may optionally introduce some redundancy intothe transmission. Any combination of the interleaver 603, mapping module604, and coder 605 may be integrated into a single module. Theinterleaver 603 simply groups the input coded bits

into N′P groups such that the mapping module 604 effectively maps eachdata bit d_(k) to a different symbol S_(p). Alternatively, a smaller orlarger number R (relative to N′P) of symbols S_(p) may be generated. Thecoding step 605 may then be adapted to convert the R input symbols S_(p)to N′P output symbols. Similarly, other combinations of the modulesshown in FIG. A2A may be consolidated. For example, the coder 602,interleaver 603, and mapping module 604 may be integrated into a singlemodule. In some cases, the function of one or more of the componentsand/or steps of the invention (and its various embodiments and aspects)may be performed by different components and/or algorithms.

The first coder 602 has a spreading factor of M. A group of K input databits d_(k) are mapped by the coder 602 to at least one set of LN′P codeddata bits

_(m). Each input data bit d₁ is mapped to M coded data bits

_(m). Each coded data bit

_(m) conveys information about one or more of the input bits d_(k). Inthis case, bit

₁ in the coded data bits 612 corresponds to bit d₁ and any M−1 previousbits. Similarly,

₂ corresponds to bits d₁ and d₂ and any M−2 previous bits. In one aspectof the invention, the last M−1 data bits of each set of K data bits maybe prepended to the set for coding purposes. This technique isparticularly useful for providing “earlier” bits relative to thebeginning of a data stream when generating a convolutional code.Alternatively, the process of prepending a cyclic prefix may begin withcyclically prepending bits to the input data bits 611 to generate alarger set of coded data symbols 612 and 613. Cyclic prefixes applied toeach subcarrier or each subchannel (a group of subcarriers) maycorrespond to signals that are functions of a predetermined number ofthe leading symbols, such as shown in the serial symbol groupings 613 or615. Typically, each cyclic prefix is a digital or analog waveformconveying the prepended bits and optionally, provided with a smoothingfunction (e.g., MSK modulation) to smooth any discontinuities betweenadjacent symbols.

In various aspects of the invention, the subcarriers shown in FIG. 7 maycorrespond to one or more users. Each user's subcarriers may bedistributed contiguously or non-contiguously. The subcarriers may bespaced equally and/or non-equally. Subcarriers may optionally correspondto different sub-space channels. In some aspects of the invention,sub-carrier sets may be distributed (e.g., spread or interleaved) overmultiple time periods to provide for time diversity. Sub-carrier setscorresponding to each user may include the same or different numbers ofsubcarriers. Each set may include equal or unequal sub-carrier spacing.Although subcarriers may be equally spaced in each set, at least twodifferent frequency spacings may be employed relative to different sets.Other variations between sets can include sub-carrier bandwidth. In oneaspect of the invention, each user's sub-carrier sets (i.e.,subchannels) may be adapted (such as relative to number of sub-carriers,channel coding, sub-carrier weighting, sub-carrier bandwidth, etc.) toprovide for adaptive sub-channel loading.

FIG. 8A is a general illustration of methods and systems that may beprovided in the present invention. Each of a set of data symbols 801 isspread such that each data symbol (or data bit) is mapped to a pluralityof coded symbols 802. Each of the coded symbols 802 are mapped to aplurality of weights 803, which then may be applied across adiversity-parameter space, such as a plurality of sub-carrierfrequencies or sub-space channels. The use of a plurality of spreadingalgorithms in place of a single large spreading algorithm enablesalgorithm, or system, reuse with the appropriate buffering and/orprocess sequencing.

The coded symbols 802 may be generated via any spreading method.Typically, channel coding (such as block coding, convolutional coding,error-check coding, etc.) spreads each data symbol over a small number Mof the coded symbols. Up to M-fold diversity is achieved when the codedsymbols are mapped onto a diversity-parameter space. If M is a smallnumber, the spreading may not provide sufficient diversity benefits.Increasing the spreading associated with channel coding may not befeasible due to the associated increase in coding complexity.Accordingly, a second spreading algorithm is appropriately adapted tospread the coded symbols 802. If the second spreading algorithm ischaracterized by a spreading rate or processing gain of O, the totalprocessing gain of both spreading operations can be up to OM.

The invention may provide for multi-level spreading operations, such asto reduce processing complexity and/or increase diversity benefits.Multi-level spreading and/or coding provides a simple architecture foradapting spreading and/or coding. Spreading algorithms of the presentinvention may employ multi-level or nested coding, such as channelcoding and/or spread-spectrum coding. More than two levels may beemployed in multi-level spreading. Furthermore, different spreadingrates may be employed relative to different channels and/or differentnumbers of subchannels. Various operations, including coding, may beperformed via parallel processing.

FIG. 8B illustrates basic components of a CI transmitter 820 and a CIreceiver 830. The transmitter 820 includes a mapping module 821, aspreading/multiplexing module 822, and a transmission module 823. Themapping module 821 is adapted to convert an input data bit stream into adata symbol stream. The mapping module 821 may further include anycombination of channel coders, spread-spectrum coders, and channelizers.The spreading/multiplexing module 822 is adapted to convert the datasymbols into a CI-coded modulated data stream.

CI coding typically spreads the modulated data stream over multiplesubchannels, such as any combination of frequency channels, time-domainsymbols, spread-spectrum codes, polarizations, sub-space channels,chirps, etc. The spreading/multiplexing module 822 may include anycombination of spread-spectrum coders, channel coders, multiple-accesscoders, multiplexers, sub-carrier generators, and invertible-transformmodules. The transmission module 823 is typically any appropriatewell-known transmitter configured to adapt an input baseband orintermediate-frequency (IF) signal for transmission into a communicationchannel 99. The transmission module 823 may include any of variousPAPR-reduction designs, such as described in the present disclosure.

A receiver module 831 coupled to the channel 99 converts receivedsignals to IF or baseband. A demultiplexer/despreader module 832processes digitized receive signals to generate estimates of thetransmitted data symbols. The module 832 may optionally include amulti-user detector 835 and/or an equalizer 836. A mapping module 833 isadapted to convert the data symbols to data bits.

The receiver module 831 typically includes any receiver adapted toconvert a received signal to a digital IF or baseband signal. Either thereceiver module 831 or module 832 may be adapted to remove any guardinterval or cyclic prefix. Module 832 is adapted to perform anynecessary demultiplexing, multiple-access processing, demodulation, anddecoding. Typically, a CI receiver will perform CI decoding. Varioustypes of equalization, interference cancellation, and/or multi-userdetection may be performed. The equalizer 836 may be adapted to performtime-domain and/or frequency-domain equalization. Various types ofinterference cancellation may be employed to mitigate co-channelinterference and jamming.

A coded CI signal that includes main and pilot signal portions ispreferably generated at the transmitter 820. The receiver 830 ispreferably adapted to estimate the frequency response of the fadingchannel using the coded pilot signals. The detected data signal and theestimated channel frequency response are used to estimate the datasignal. The determination can be based on a channel inversion of thefrequency response or a new channel estimation combined with maximumlikelihood sub-sequence estimation. Other techniques for channelestimation and signal detection for multi-carrier systems may beemployed that do not necessarily rely on coded pilot signals.

The invention may utilize an iterative maximum likelihood (ML)estimation method and system to estimate the impulse response of amultipath fading channel and to detect a transmitted signal in OFDM andother multicarrier systems. A system and method may be employed thatiteratively finds the joint channel impulse response and the transmittedsignal to maximize the likelihood of estimating the correct main signal.The estimation procedure can start with a CI symbol with pilot signals.In this case, an initial maximum likelihood estimate of the impulseresponse of the channel is obtained from the pilot signals. Based on theinitial estimate of the impulse response, a first estimation of the mainsignal is made. After the initial estimate, both the pilot signals andthe estimated main signals are fed back to the channel estimation stepto obtain an improved estimation of the channel's impulse response.Then, an updated estimation of the main signal can be obtained using there-estimated channel impulse response. The iteration procedure stopswhen improvement on the channel estimation is below a predeterminedthreshold.

For CI signals that do not include pilot signals, the iteration startsby assigning the initial estimation of the impulse response of thechannel to be that of the final estimation of the previous CI symbol ordecoded data symbol. The other iteration procedures follow those stepsdescribed in the previous paragraph.

The multi-user detector 835 may be adapted to cancel ISI and/or MAI asappropriate. The multi-user detector 836 may provide a hard decision ora soft decision output to a decoder in module 832. In some cases, it isnecessary for the multi-user detector 836 to generate a soft decision bycombining a hard-decision output with a confidence measurement, such asmay be provided during multi-user detection and/or measured with respectto some predetermined expected signal levels. The multi-user detector836 may be adapted specifically to compensate for loss in orthogonalitybetween symbols (or users) for sub-optimal equalization, such as whenthe received signal is distorted by deep fades and/or high levels ofinterference.

The mapping module 833 typically decodes (e.g., maps) the data symbolsinto corresponding bit values. Various types of channel decoding may beemployed, as required. The mapping module may perform iterative feedbackoperations that may include either or both hard decision and softdecision processing.

FIG. 8C illustrates a receiver system and method of the inventionadapted to process received FD-CI signals. A receiver module 851processes received multicarrier signals and generates a down-converteddigital signal corresponding to at least one user. A cyclic prefixremoval module 852 discards any cyclic prefix or guard intervalprepended to the received signal. A filter bank, such as a DFT 853,decomposes the signal into a predetermined set of spectral components. Afrequency-domain equalizer (FEQ) 854 is adapted to compensate forchannel distortion. The FEQ 854 typically employs some form of channelestimation. A sub-channel demultiplexer 855 processes the equalizedsub-channels to produce a plurality of spread data symbol values. Thedemultiplexer 855 may be adapted to combine each of a plurality ofsub-carrier sets or values. A despreader 856 is adapted to provideappropriate de-spreading codes to spread data symbols to produce aplurality of estimated data symbol values. An optional MUD 857 may beused to compensate for ISI, such as due to loss of one or more of thesubcarriers. A symbol-mapping module 858 is adapted to map the datasymbol values to a set of data bits or a second set of data symbols. Thesymbol-mapping module 858 may include a decoder. Optionally, a separatedecoder 859 is provided, such as to perform bit-level decoding.

FIG. 9A illustrates a distribution of sub-carrier weights w_(n) over aplurality of frequency-time channels. In particular, the sub-carrierweights w_(n) are interleaved in time, such as to further reduce impulsenoise or compensate for quickly varying channel and/or interferenceconditions. In FIG. 9B, weights are applied to time-frequency channelsin a digital-chirp (i.e., frequency ramp) format. In FIG. 9C,sub-carrier weights may share the same time slots.

The invention can provide for tone hopping on transmitter arrays whereineach transmitter element of the array is provided a subsequentsub-carrier frequency corresponding to a predetermined hopping pattern(e.g., a FHSS code) and hop interval. Multiple carriers transmitted byeach element of a transmitter array may be hopped (i.e., provided with asubsequent set of carrier frequencies).

FIG. 10A illustrates a CI transmitter adapted to reduce PAPR ofmulticarrier signals. A CI symbol generator 301 processes an input databit stream d_(k) to generate a sequence of symbols S_(n) characterizedby CI coding. Since CI coding may employ code chips that are similar toorthogonal basis terms in a Fourier transform, a fast algorithm similarto an FFT may be employed for mapping data symbols onto CI codes. The CIsymbol generator 301 may employ channel coding, spreading, and/ormultiple-access coding.

A sub-carrier generator 302 is adapted to map each of the N symbolsS_(n) onto a plurality of carriers that typically equals N. The numberof camers may be greater or less than the number N of symbols S_(n). Thesub-carrier generator 302 may include an IDFT, an IFFT, a pulsegenerator, or any other suitable form of sub-carrier generator. e.g.,302.1 to 302.N.

The output coded subcarriers are combined in a combiner 303 including aplurality M of combiner modules 303.1 to 303.M. The combiner modules303.1 to 303.M may each be adapted to combine a plurality of equallyspaced subcarriers. Different numbers of subcarriers may be combined byeach combiner module 303.1 to 303.M. In one embodiment of the invention,contiguous subcarriers between unused tones or pilot tones are combinedby each of the combiner modules 303.1 to 303.M. In another embodiment ofthe invention, equally spaced subcarriers distributed over multiplefrequency bands are combined in each of the combiner modules 303.1 to303.M.

Signal outputs from the combiner modules 303.1 to 303.M arecharacterized by low PAPR due to the CI encoding. Even when amplitudeshift-key modulation is employed, the present invention can reduce theeffects of PAPR at the power amplifier by synthesizing high-PAPRconstellation symbols from combinations of low-PAPR components.Additional PAPR-reduction techniques may be employed, as described inthe present disclosure. The signal outputs are provided with a guardinterval or cyclic prefix by a guard interval module 304. Theguard-interval module 304 may optionally include a plurality ofguard-interval modules 304.1 to 304.M. A guard interval corresponding toeach combined signal may be generated.

Each of the combined signals is provided to a transmitter module 305. Insome cases, CI coding or some other PAPR-reduction process may beapplied to the combined signals to ensure low PAPR of the signals priorto amplification. Alternatively, the transmitter module 305 may includea plurality of transmitter modules 305.1 to 305.M adapted to processeach of the combined signals. In one aspect of the invention, eachtransmitter module 305.1 to 305.M includes a power amplifier adapted toseparately amplify each of the combined signals. The amplified signalsmay then be combined in a coupling system, such as an antenna (notshown), a waveguide (not shown), or a multi-port junction (not shown).The transmitter modules 305.1 to 305.M may be provided with separateantennas, or else the transmitter modules 305.1 to 305.M may share oneor more antennas.

FIG. 10B illustrates an embodiment of a CI symbol generator 301. Inputdata symbols d_(k) are mapped 308 to symbol constellations d_(k), priorto a CI code transform module 309. The symbol mapping module 308 mayprovide channel coding, code division multiple access, andlorspread-spectrum coding. The CI code transform module 309 can implementorthogonal basis functions of any invertible transform and provide forany appropriate fast transforms. Similarly, a receiver employing CIdecoding may implement a fast transform.

FIG. 10C illustrates an alternative embodiment of a CI transmitter. Amulti-carrier generator 311 provides for sub-carrier selection. Acombiner 312 including a plurality of combiner modules 312.1 to 312.Mcombines subsets of the subcarriers to produce a plurality M of CIwaveforms. Alternative methods and systems may be used to generate CIwaveforms, such as time-domain coding techniques disclosed in Applicantspending patent application entitled “Time-Domain Applications of BasicCarrier Interferometry Codes for Spectrum Allocation,” which isincorporated by reference herein.

A modulator 313 is adapted to impress a vector of data symbols onto theCI waveforms. The modulated signal outputs are provided with a guardinterval or cyclic prefix by a guard interval module 314. Theguard-interval module 314 may optionally include a plurality ofguard-interval modules 314.1 to 314.M. Each of the combined signals isprovided to a transmitter module 315. The transmitter module 315 mayinclude a plurality of transmitter modules 315.1 to 315.M adapted toprocess each of the combined signals.

FIG. 11A illustrates a CI pulse train consisting of a number N of pulses1111 to 1119 generated from a superposition of N equally spacedsub-carrier frequencies f_(n)=f₁ to f_(N). The symbol duration T_(s)equals the inverse of the subcarrier frequency spacing f_(s). The pulses1111 to 1119 are orthogonally spaced in time. That is, the pulses 1111to 1119 are centered at N instants in time uniformly spaced over thesymbol interval T_(s). Orthogonal positioning of CI pulses is well knownin the art, such as described in PCT Patent Application PCT/US99/02838,U.S. Pat. No. 5,955,992, and C. R. Nassar, B. Natarajan, and S. Shattil,“Introduction of carrier interference to spread spectrum multipleaccess”, IEEE Emerging Technologies Symposium, Dallas, Tex., Apr. 12-13,1999, which are incorporated by reference.

Each pulse in FIG. 11A corresponds to a set of phase offsets (i.e., anorthogonal CI code) applied to the component subcarriers. For example, afirst set of codes 1111 applied to the subcarriers produces asuperposition of the carriers to produce a first pulse 1101 centered ata first time instant t=0. Similarly, codes 1112 to 1119 generate CIpulses 1102 to 1109, respectively. The generation of CI codes is wellknown in the art and described in the previously cited publications onCI.

FIG. 11B illustrates a set of decoding symbols 1120 applied tosub-carrier frequencies f_(n)=f₁ to f_(N). The decode set 1120 typicallyincludes a complex conjugate of at least one of the codes 1111 to 1119.The decode set 1120 is applied to measured sub-carrier weights torecover data symbols impressed onto at least one of the correspondingpulses 1111 to 1119.

CI coding is the process of distributing coded data across multiplesub-carrier frequencies to generate a superposition pulse substantiallyconfined to a narrow pulse interval. CI coding is an invertibletransform. Thus, one aspect of the invention recognizes that CI codingmay equivalently be applied across multiple CI pulses (or other pulsewaveforms) in a symbol interval to isolate or select one or moresub-carrier components.

FIG. 11C illustrates a set of decoding symbols 1120 applied to CIpulses, which are represented by CI codes 1111 to 1119. The decodingsymbols 1120 are applied to the pulse train spanning a symbol durationT_(s). The resulting waveform corresponds to at least one of thesub-carrier components. For example, if the decoding-symbol set 1120 isthe complex conjugate of one of the CI codes 1111 to 1119, the resultingwaveform is one of the CI sub-carrier frequencies.

FIG. 11D illustrates an exemplary method of the invention. A set ofuniformly spaced pulses 1101 to 1109 (such as may be generated via CIpulse generation, or some other means) is provided. Desired spectralcharacteristics may be established by controlling one or more signalingparameters, such as pulse repetition period, number of pulses, and pulseshape.

In a method of the invention, one or more CI codes 1120, includingaggregate CI codes (i.e., linear combinations of basic CI codes), may beapplied to the pulses 1101 to 1109. Each CI code chip may multiply, ormodulate, a corresponding pulse 1101 to 1109. The resulting time-domainsignal is composed of one or more of the pulse signal components (i.e.,sub-carrier frequencies) depending on the code 1120. In the event thatthe code 1120 is an aggregate code, a plurality of orthogonal signalsmay be generated by cyclically shifting copies of the generatedtime-domain signal. The appropriate time offsets may be selected basedon the number of sub-carrier components, the number of desiredorthogonal signals (the maximum number equals the number of components),and/or via sliding correlation. Data symbols 1122 may be modulated ontothe resulting time-domain signals or impressed onto the code 1120.

The present invention provides a CI-type modulation scheme that enablesdifferent users in a communication system to transmit simultaneously atdifferent data rates (or different effective bandwidths) while providingfrequency diversity, low PAPR, and orthogonality between the users.Additionally, the improved modulation scheme permits transmitters withinthe system to vary their data rates (or effective bandwidths), adapt tochanging spectral requirements, and/or adapt to changing channelconditions while preserving low PAPR and orthogonality.

FIG. 12 is a block diagram of an exemplary system 1200 in accordancewith the present invention. The system 1200 includes one or moretransmitters 1202, each corresponding to a particular user andcommunicating with one or more receivers (not shown) by way of acommunication channel 99. The communication channel 99 can be a radiofrequency (RF) channel in a multi-user communication system, such as acellular system, wireless network, mobile radio system, or the like.

Multiple users can simultaneously transmit data over the channel 99. Thetransmitters 1202 and receivers (not shown) can be fixed or mobilesubscriber units and/or base stations. The transmitters 1202 andreceivers (not shown) can include suitable combinations of hardwareand/or software components for implementing the modulation scheme of thepresent invention. As shown, the transmitter 1202 illustrated in FIG. 12includes a symbol-to-time mapping module 1204, a pulse filter 1206supported by a plurality of sub-modules (such as a pulse-train generator1205, a CI code generator 1203, and a carrier selector 1201) a cyclicextension device 1210, an optional pulse-shaping filter 1212, and atransmitter module 1214.

The system 1200 is described further herein for the case where userswithin the CI-type system can have different data rates. Various codingtechniques for achieving orthogonal discrimination for different datarates and/or sub-carrier allocations are disclosed. The conditions formaintaining orthogonality, i.e., minimal multiple access interference(MAI), between the different users are also described.

In the subject version of CI signaling, a user's baseband signal beginsas a single-carrier phase shift keying (PSK) or quadrature amplitudemodulation (QAM) symbol stream. The symbols and/or the data bits thatmake up the symbols may optionally be provided with coding. The symbolsare mapped 1204 to a number of pulse positions per symbol durationcorresponding to the number of subcarriers allocated to the user. Thesymbols are impressed onto time-domain CI waveforms via coded CI pulsefiltering 1206. The pulse filter 1206 either applies the time-domainwaveform to each mapped symbol, or the pulse filter centers a pluralityof time-domain waveforms at the predetermined pulse positions andmodulates each waveform with at least one of the data symbols.

The pulse filter 1206 may provide time-domain waveforms generated by apulse-train generator 1205, a CI coder 1203 adapted to apply one or moreCI codes to the pulse train, and a carrier selector 1201 adapted todirect code selection by the CI coder 1203. The repetition of the pulsesproduced by the pulse-train generator 1205 causes the spectrum of pulsetrain to be non-zero only at certain sub-carrier frequencies. The codesapplied to the pulse train by the CI coder 1203 select a predeterminedset of the sub-carrier frequencies.

The pulse filter 1206 may include memory to store time-domain waveforms.The pulse filter 1206 may include signal replicators, delay systems,interleaver, and/or any other appropriate analog or digital componentsnecessary for the generation and/or application of a plurality oftime-domain waveforms. The pulse filter 1206 may be adapted to overlayand/or interleave a plurality of modulated time-domain waveforms. Thepulse filter 1206 may include a CI pulse-shaping filter, a root-raisedcosine filter, an MSK filter, or the like. In another aspect of theinvention, the CI coder 1203 may be adapted to provide data-modulated CIcodes to the pulse train.

A guard interval or cyclic prefix may optionally be provided to thetime-domain signal. The cyclic extension device 1210 may optionally bepositioned between the symbol-to-time mapping module 1204 and the pulsefilter 1206 or between the optional pulse-shaping filter 1212 and thetransmitter module 1214. Optional pulse shaping 1212 may be used toprovide the spectrum with a steeper roll off compared to OFDM. Thetransmitter module 1214 may optionally include filters, such as low-passfilters and/or anti-aliasing filters.

The CI transmissions remain orthogonal as long as: 1) they occupydifferent sets of subcarriers, which is accomplished by user-specific CIcodes, 2) a cyclic extension (or guard period) is added to thetransmission to compensate for any channel delay spread, and 3) thesignals are synchronized with the receiver in time and frequency.

FIG. 13 shows a flowchart illustrating one possible variation of theoperation of the transmitter shown in FIG. 12. For each active user in acommunication system, a predetermined set of at least one possible setof subcarriers is selected 1301. Carrier selection 1301 may be adaptiveand/or may be in response to measured (and/or estimated) channelconditions, transmission throughput requirements, sub-carrierallocations, etc. Symbol mapping 1302 establishes a number oftransmission symbols per symbol interval. Data symbols and/ortime-domain waveforms are mapped to predetermined instants in time(i.e., phase spaces) dictated by any combination of the number ofsubcarriers and the time-domain waveform(s) employed by the transmitter.

Pulse filtering 1303 maps one or more time-domain waveforms (which havea frequency spectrum corresponding to the carrier selection) to thephase spaces. Pulse filtering 303 may include modulating data symbolsonto the waveforms, or equivalently, mapping the waveforms onto symbolsallocated to the phase spaces. In exemplary embodiments of theinvention, pulse filtering 1303 includes applying time-domain coding1311 based on CI to a train of pulses. The pulses may be generated fromsuperpositions of subcarriers, or they may be generated usingtime-domain generation and shaping techniques. Pulse filtering 1303 mayinclude a parallel process of impressing data symbols onto time-shifted(e.g., cyclically shifted) waveforms, followed by combining (e.g.,summing) the waveforms.

In some applications in which time-domain waveforms include gaps (orzeros), the process of combining different data-modulated waveforms maybe implemented as an interleaving process. Such combining processes maybe performed as a serial process. Accordingly, carrier selections mayadvantageously be made to provide time-domain waveforms with gaps suchthat combining the time-domain waveforms may be implemented viainterleaving and/or other appropriate serial processing.

In step 1304, a guard period (a cyclic extension comprising a prefix,postfix, or both) may be provided prior to, and/or following, pulsefiltering 1303. The cyclic prefix step 1304 may be provided before orafter pulse shaping 1305. A transmit step 1306 processes the signal fortransmission into a communication channel 99.

In step 1303, a user-specific modulation code is applied. The modulationcode can be any suitable code meeting the code-assignment conditionsdisclosed herein. With proper selection of modulation codes,orthogonality between the different users can be maintained, even inmultipath conditions, as long as the channel varies slowly with respectto the symbol duration T_(s). To maintain orthogonality, the symbolduration can be the same for all users. Similarly, different datastreams, and/or data symbols corresponding to a particular user may beconveyed with a plurality of different user codes. For example, codeddata symbols may be spread over multiple sub-carrier channels in a waythat simultaneously achieves enhanced frequency diversity for each databit and the reduced complexity of frequency-division processing.

The present invention provides for multi-rate transmissions by usinguser-specific data block and repetition sizes, as well as user-specificmodulation codes, thereby providing different data rates to differentusers and hence a high degree of data rate flexibility. It alsopreserves a low PAPR. In some applications of the invention, the effectsof PAPR may be further reduced by separately amplifying (and optionally,providing for separate transmission) a plurality of the modulatedtime-domain waveforms and/or sets of waveforms.

Variations to the systems and methods of the invention may be made inaccordance with the prior art without departing from the scope of theinvention. Similarly, other aspects of related forms of communication(e.g., multi-carrier and/or single-carrier communications) may becombined with the present invention. Furthermore, the order of thevarious illustrated components and steps may be changed withoutdeparting from the scope of the invention.

FIG. 14 shows a first receiver structure in which channel equalizationis performed prior to correlating the signal with a user-specificmodulation code. The first receiver includes an RF circuit 1402 adaptedto perform RF-to-baseband conversion. The RF circuit 1402 may include anA/D converter sampling circuit (not shown), a baseband filter 1404, acyclic extension remover 1406, an equalizer 1408, a decoder 1410, and asymbol decision device 1412. The decoder 1410 and symbol decision device1412 perform their functions for specific users.

The received signal is down converted to a complex baseband signal,filtered (image rejection, adjacent channel rejection, avoid aliasing),and digitized by the RF circuit 1402. The baseband signal may bebaseband filtered, such as by a filter 1404 matched to the transmitpulse-shaping filter. The cyclic prefix remover 1406 removes any cyclicextension or guard interval. The frequency domain equalization module1408 equalizes the received signals for each user's (corresponding tothe number of user signals the receiver is adapted to process) channelresponse. As each user is allocated a unique set of orthogonalsubcarriers (due to the modulation codes), all users can be equalizedsimultaneously in the frequency domain using only one N-point transform.However, the equalizer coefficients are different for each user and areapplied only to the sub-carriers occupied by that user. Thus theequalizer 1408 may perform a combination of common processing anduser-specific processing. Alternatively the equalization can be based onother techniques such as linear transversal time-domain equalization,decision feedback equalization, maximum likelihood sequence estimation,iterative equalization, inter-symbol-interference (ISI) cancellation,andlor turbo equalization.

For each user, the equalized signal is then code correlated by the CIdecoder 1410. Channel decoding, such as Forward Error Correction (FEC)decoding may be provided if error correction coding was used in thetransmitter. A decision device 1412 may be adapted to perform a logicdecision based on the estimated symbols to determine the values of thesymbols.

Variations to the receiver processor may be provided without departingfrom the spirit and scope of the invention. For example, in analternative receiver architecture, the correlation with user-specificcodes is performed prior to channel equalization.

In various designs of the receiver, time-domain codes (such as CI codesillustrated herein) may be applied to a received time-domain signal toextract predetermined sub-carrier values. Thus, in the same way thattime-domain CI coding may be used to map data symbols onto individualsub-carrier frequencies (thus, eliminating the need for typical IFFTprocesses at the transmitter), CI decoding may be provided in place ofany FFT process, such as typically performed at the receiver.

The preceding descriptions merely illustrates the principles of theinvention. It will thus be appreciated that those skilled in the artwill be able to devise various arrangements which, although notexplicitly described or shown herein, embody the principles of theinvention and are included within its spirit and scope. Furthermore, allexamples and conditional language recited herein are principallyintended expressly to be only for pedagogical purposes to aid the readerin understanding the principles of the invention and the conceptscontributed by the inventor(s) to furthering the art, and are to beconstrued as being without limitation to such specifically recitedexamples and conditions. Moreover, all statements herein recitingprinciples, aspects, and embodiments of the invention, as well asspecific examples thereof are intended to encompass both structural andfunctional equivalents thereof. Additionally, it is intended that suchequivalents include both currently known equivalents as well asequivalents developed in the future, i.e., any elements developed thatperform the same function, regardless of structure.

Thus, for example, it will be appreciated by those skilled in the artthat the block diagrams herein represent conceptual views ofillustrative circuitry embodying the principles of the invention.Similarly, it will be appreciated that any flow charts, flow diagrams,state transition diagrams, codes, and the like represent variousprocesses which may be substantially represented in computer readablemedium and so executed by a computer or processor, whether or not suchcomputer or processor is explicitly shown.

The functions of the various elements shown in the drawings, includingfunctional blocks labeled as “processors,” may be provided through theuse of dedicated hardware as well as hardware capable of executingsoftware in association with appropriate software. When provided by aprocessor, the functions may be provided by a single dedicatedprocessor, by a single shared processor, or by a plurality of individualprocessors, some of which may be shared. Moreover, explicit use of theterm “processor” or “controller” should not be construed to referexclusively to hardware capable of executing software, and mayimplicitly include, without limitation, digital signal processor (DSP)hardware, read-only memory (ROM) for storing software, random accessmemory (RAM), and non-volatile storage. Other hardware, conventionaland/or custom, may also be included. Similarly, any switches shown inthe FIGS. are conceptual only. Their function may be carried out throughthe operation of program logic, through dedicated logic, through theinteraction of program control and dedicated logic, or even manually,the particular technique being selectable by the implementer as morespecifically understood from the context.

In the description hereof any element expressed as a means forperforming a specified function is intended to encompass any way ofperforming that function including, for example, a) a combination ofcircuit elements which performs that function or b) software in anyform, including, therefore, firmware, microcode or the like, combinedwith appropriate circuitry for executing that software to perform thefunction. The invention as defined herein resides in the fact that thefunctionalities provided by the various recited means are combined andbrought together in the manner which the operational descriptions callfor. Applicant thus regards any means which can provide thosefunctionalities as equivalent as those shown herein.

1. A carrier interferometry (CI) transmission system employingpeak-to-average power ratio (PAPR)-reduction signaling, the CItransmission system including: a symbol-mapping module adapted toallocate a predetermined number of data bits to a predetermined set ofsubchannels, a CI coder adapted to perform at least one predeterminedcombination of data spreading and channel coding to produce a pluralityof input symbols, a carrier-generator module adapted to associate theinput symbols with at least one set of subchannels and generate acorresponding time-domain sequence representing a data-payload signal,and an unloaded channel-encoding module adapted to select unloadedsubchannels for transmission of at least one PAPR-reduction signal. 2.The CI transmission system recited in claim 1 wherein the unloadedchannel-encoding module is adapted to select and generate at least oneunloaded subchannel for combining with the time-domain sequence when thetime-domain sequence exceeds a predetermined power threshold.
 3. The CItransmission system recited in claim 2 wherein the unloadedchannel-encoding module is adapted to generate PAPR-reduction signals inunloaded subchannels and combine the PAPR-reduction signals with thetime-domain sequence until the time-domain sequence power drops below apredetermined threshold.
 4. The CI transmission system recited in claim1 wherein the symbol-mapping module is adapted to generate unloadedsubchannels by not loading subchannels that are comprised by adversechannel conditions.
 5. The CI transmission system recited in claim 1wherein the unloaded channel-encoding module is adapted to maintain thedata-payload signal below a predetermined clipping threshold.
 6. The CItransmission system recited in claim 1 wherein the unloadedchannel-encoding module is adapted to combine the at least onePAPR-reduction signal with at least one of the plurality of inputsymbols and the data-payload signal.
 7. The CI transmission systemrecited in claim 1 wherein the symbol-mapping module is adapted to ceaseloading at least one predetermined subchannel that is below at least onepredetermined channel-quality metric such that the unloadedchannel-encoding module is capable of selecting said predeterminedsubchannel for transmission of at least one PAPR-reduction signal. 8.The CI transmission system recited in claim 1 wherein the symbol-mappingmodule is adapted to allocate a predetermined number of data bits to atleast one of a set of subchannels including space-frequency subchannels,space-time subchannels, CI phase-space subchannels, spatialsub-channels, and polarization subchannels.
 9. The CI transmissionsystem recited in claim 1 wherein the symbol-mapping module is furtheradapted to select which of a plurality of sequence permutations of thepredetermined number of data bits results in the greatest reduction ofPAPR in the data-payload signal.
 10. A multicarrier transmission systemadapted to reduce the effects of high peak-to-average power ratio (PAPR)including: a carrier interferometry (CI) coder adapted to spread atleast one data sequence with at least one set of CI codes for generatingat least one set of CI-coded symbols, a sub-carrier generator adapted tomap the at least one set of CI-coded symbols onto a plurality ofsubcarriers, a plurality of combiners adapted to combine sets of theplurality of carriers for producing a plurality of CI-coded time-domainsequences that are characterized by low PAPR, and a plurality of poweramplifiers coupled to the plurality of combiners, the amplifiers adaptedto amplify the plurality of CI-coded time-domain sequences.
 11. Themulticarrier transmission system recited in claim 10 further includingan amplified-signal combiner coupled to the plurality of poweramplifiers.
 12. The multicarrier transmission system recited in claim 11wherein the amplified-signal combiner includes at least one of a setincluding an antenna, a waveguide, and a multi-port junction.
 13. Amulticarrier signal generator including: a pulse-train generator adaptedto generate a sequence of pulse waveforms having a predeterminedspectrum, a carrier interferometry (CI) coder capable of generating atleast one CI code, and a carrier selector coupled to the CI coder andthe pulse-train generator, the carrier selector adapted to impress theat least one CI code onto the sequence of pulse waveforms to shape thepredetermined spectrum.